A typical method of controlling the current of a voltage inverter is proportional integral control of the dq axis current. FIG. 1 is a layout diagram showing a prior art example of an AC motor control device using such current control. In FIG. 1, a voltage inverter 3 inputs a voltage that is smoothed by a smoothing capacitor 2 that smoothes the DC voltage from a DC power source 1, converts the DC power from this DC power source 1 to 3-phase AC power and supplies this as drive power to a motor 4. The motor current of the motor 4 is detected by Hall CTs 5U, 5V and 5W and input to a current detection circuit 13. The current detection circuit 13 outputs the output signals of the Hall CTs 5U, 5V and 5W to a co-ordinate conversion circuit 14 as detected currents iu, iv, iw in accordance with the scaling in the control circuit. Also, the rotor position of the motor 4 is detected by a rotation sensor 6 and input to a rotation detection circuit 7. The rotation detection circuit 7 finds and outputs an electrical angle signal θr and speed or corresponding to the position of the rotor from the output signal of the rotation sensor 6.
The deviation of the speed or detected by the rotation detector 7 with respect to a speed reference or* is found by a subtractor 8 and this speed deviation is input to a speed control circuit 9. The speed control circuit 9 amplifies the speed deviation that is output by the subtractor 8 and adjusts a torque instruction Trq* so that the speed or tracks the speed reference or*.
A flux weakening function generator 10 inputs a reinforcing flux reference Φ** and the speed or; below a prescribed speed, the flux weakening function generator 10 outputs the reinforcing flux reference Φ** without modification, but, above the prescribed speed, it outputs the reinforcing flux reference Φ** as a flux reference Φ* that is weakened in inverse proportion to the speed. A vector calculation circuit 11 calculates and outputs a torque-based current reference iq*, flux-based current reference id* and slip angle θs based on the flux reference Φ* and torque reference Trq*. An adder 12 adds the rotor position signal θr from the rotation detection circuit 7 and the slip angle θs* from the vector calculation circuit 11 and outputs a flux position signal θo to coordinate conversion circuits 14, 17. The co-ordinate conversion circuit 14 converts the detected currents iu, iv and iw from the current detection circuit 13 to a flux-based detected current id and torque-based detected current iq on the dq axis co-ordinates synchronized with the flux of the motor 4, by using the flux position signal θo from the adder 12.
Next, a subtractor 15d calculates the deviation of the flux-based detected current id from the co-ordinate converter 14 and the current reference id* of the d axis from the vector calculation circuit 11, and a subtractor 15q calculates the deviation of the torque-based detected current iq from the co-ordinate converter 14 and iq* of the q axis from the vector calculation circuit 11 and outputs these respectively to current control circuits 16d and 16q. The current control circuits 16d, 16q perform proportional integration and amplification on the current deviations that are output by the subtractors 15d, 15q and output the results to the co-ordinate conversion circuit 17 as voltage instructions vd*, and vq*. The co-ordinate conversion circuit 17 converts the voltage instructions vd*, vq* to voltage instructions vu*, vv*, vw* of a stator static co-ordinate system using the flux position signal θo and outputs these to a PWM control circuit 18. The PWM control circuit 18 delivers output to an inverter 3 that outputs pulse trains whose duty varies in accordance with the respective magnitudes of the voltage instructions vu*, vv* and vw*.
In the case of the AC motor control device shown in FIG. 1, the 3-phase AC currents iu, iv and iw that were detected are converted by the co-ordinate converter 14 to DC quantities id, iq on the dq axis co-ordinates based on the flux phase of the motor, and the deviations (id*-id), (iq*-iq) with regard to the respective current references are amplified by the proportional integration type current control circuits 16d, 16q. Voltage instructions vd*, vq* are then found in accordance with these amplified quantities and converted by the co-ordinate conversion circuit 17 to the stator static co-ordinates-based voltage instructions vu*, vv* and vw* using the flux position signal θo: these are then supplied as voltage references to the PWM control circuit 18 for, for example, triangular wave comparison PWM (pulse width modulation) and inverter control is performed with the PWM signal that is output by this PWM control circuit 18. That is, since the current is subjected to proportional integral control by conversion to DC quantities on the dq axis coordinates, control without steady deviation (or steady-state deviation) can be achieved even in the case of high-frequency AC current of frequency as high as some hundreds of Hz.
In current control systems for motors, control devices for voltage type inverters are available whereby the change of voltage on switching of pulse number in the case of high rotational speed/few pulses can be reduced compared with conventionally, by making the size of current ripples more uniform and smaller, and by reducing back pulses in comparison with conventionally. An example is Laid-open Japanese Patent Application No. 2003-235270 (Patent reference 1).
There are also available inverter control devices in which switching control is performed whereby PWM control is realized that makes possible PWM control with few harmonics in a steady condition and high-speed current control in a transitory condition. An example is Japanese Patent No. 3267524 (Patent reference 2).
Hereinbelow, we shall use the term current tracking PWM to refer to a PWM system of the current tracking type that generates a direct PWM signal such that the detected current tracks a current reference, as in Patent Reference 2.
FIG. 2 is a layout diagram (or a block diagram) showing another prior art example of an AC motor control device employing current control. The example of FIG. 2 is an example in which the dq axis current control section and the PWM control circuit 18 of the device of FIG. 1 are replaced by current tracking type PWM. In FIG. 2, the torque-based current reference iq* and flux-based current reference id* that are output from the vector calculation circuit 11 are converted to the stator static co-ordinate 3-phase current references iu*, iv* and iw* by the co-ordinate conversion circuit 19, their respective differences from the 3-phase detected currents iu, iv, iw obtained by the subtractors 20U, 20V and 20W and then supplied to the current tracking type PWM control circuit 21.
The current tracking type PWM control circuit 21 generates PWM signals such that the detected currents iu, iv, iw track the current references iu*, iv* and iw* and these PWM signals perform on/off control of the constituent switching elements of the inverter 3. With this system, no carrier wave is generated and the current response is extremely fast, since PWM signals are directly generated such that the current tracks the instruction values.
However, in the prior art example shown in FIG. 1, the current control response is affected by the modulation frequency of downstream PWM control. Also, if the integrator output on the q axis side exceeds the level corresponding to the q axis voltage that is actually capable of being output, the system falls into a condition in which current control cannot be achieved and, so, in the high-speed region, it is necessary to weaken the flux rather earlier: thus the output capacity of the motor is restricted and operating efficiency is lowered.
Also, in the case of the PWM control circuit 18, it is necessary to change over the PWM control system in accordance with the operating frequency of the motor. Specifically, in the range in which the operating frequency of the motor is low, asynchronous PWM is performed in which PWM signals are generated by comparing a triangular carrier wave of fixed modulation frequency and a voltage reference sine wave; however, when the operating frequency becomes high, approaching the frequency of the voltage reference sine wave and triangular carrier, fluctuation of the fundamental wave component contained in the PWM signal becomes large, so synchronous PWM is performed wherein voltage fluctuation is eliminated by maintaining the frequency of the triangular carrier wave at an integer multiple of the voltage reference sine wave. Furthermore, when the operating frequency becomes high, in the region in which operation is conducted with an extremely low number of pulses of the PWM signal per cycle of the operating frequency, such as for example 5 pulses or 3 pulses per cycle of the operating frequency, PWM is performed in accordance with a pulse pattern such as to preferentially remove low order harmonics such as fifth- or seventh-order harmonics, which have a large effect on the efficiency of the motor.
Now, in control combining dq axis current control and PWM control, the current control lags, so it is not possible to use current control to suppress voltage fluctuations arising from the low order harmonic voltages that arise from PWM control, or arising from frequency differences of the carrier wave and the voltage reference. It is therefore necessary to perform PWM control in such a way that the PWM control circuit 18 does not output PWM signals such as to produce undesirable harmonic voltages or voltage fluctuations.
However, in the event of changeover of the PWM control method, the output voltage changes and torque fluctuation is generated by the rapid change of current produced by this voltage change: in severe cases, the overcurrent protection system may be actuated. It is therefore necessary to effect changeover by selecting the phase such that abrupt current changes are not produced; however, during this changeover, transitional changeover control is necessary such as restriction of the current references. Such adjustment to restrict the current references is troublesome and, depending on the application, it is sometimes not possible to adopt a changeover system involving restriction of the current references.
In the case of the prior art example shown in FIG. 2, the current response is extremely fast, and the current control response is not limited by the modulation frequency as it is in the case of the system of FIG. 1. Also, since the PWM waveform is automatically and continuously changed over in accordance with the operating frequency, there is no need for deliberate changeover of the PWM control. Furthermore it is possible to shift continuously to single pulse operation without falling into a condition in which control is impossible in the high-speed region.
However, a characteristic drawback of current tracking PWM is the existence of a theoretically steady error (or steady-state error). Since, in current tracking PWM, the PWM signal is generated in a magnitude relationship in comparison with the instantaneous value, the proportional gain is infinitely large. Since if this PWM signal is directly used for operational purposes, the frequency of the PWM signal is too high, an insensitive zone provided by hysteresis or a delay time imposed by a timer is provided: however, a steady error is produced by such an insensitive zone or delay time. If the switching frequency is high, the steady error is small, but if the switching frequency is low the steady error increases and has a considerable effect on the performance of the motor.
A considerable merit of current tracking PWM is that high-speed response is obtained irrespective of the switching frequency. Large drives for industrial use and main motor drives for electric vehicles etc employ large-current switching elements, so switching losses are considerable. Consequently, the minimum switching frequency is adopted at which the necessary current response can be obtained, in order to moderately satisfy both performance and efficiency. Employment of current tracking PWM in such applications makes it possible to enormously improve performance, since the current response can be speeded up without needing to raise the switching frequency. Indeed, consideration may be given to improving both the performance and efficiency while positively lowering the switching frequency.
Next, FIG. 4 is a characteristic diagram showing the difference in voltage output capability of an inverter depending on the control system employed and the corresponding change of the flux weakening control region. In the example of FIG. 1, sine wave PWM control is performed by PWM control circuits 16d, 16q downstream of a proportional integral current control circuit 18. If the DC voltage of the inverter is assumed to be Edc, the maximum value of the line voltage (instantaneous value) that can be output by the voltage type inverter 3 is ±Edc. The maximum sine wave voltage is therefore |±Edc·sin θ|. This is the curve S0 in FIG. 4 (theoretical limit in the case of sine wave PWM), the flux instruction being the curve S0′ (theoretical limit in the case of sine wave PWM).
However, as described above, if the output (voltage instruction) of the current control circuit 16q exceeds the voltage output capability, control becomes impossible. Consequently, in order to provide a voltage margin, a voltage instruction (saturation level of the current controller 16q) of for example 95% of the voltage output capability in sine wave PWM is employed, as indicated by the curve S1 (practical limit in the case of sine wave PWM) in the upper part of FIG. 4. The flux instruction in this case is the lower curve S1′ (practical limit in the case of sine wave PWM) of FIG. 4.
The maximum voltage of a PWM inverter need not be a sine wave: at any rate, if the maximum is desired, this can be achieved without using PWM at all by outputting a square wave voltage, achieved by obtaining output in which for an electrical angle of 180° the positive side elements of an inverter 3 are turned ON and for the remaining 180° in the electrical angle the negative elements are turned ON. This mode will hereinbelow be referred to as single pulse mode. The magnitude of the fundamental component of the output line voltage is then expressed by ±(2√{square root over (3/π)})·Edc·sin θ, the magnitude of the amplitude being 1.103·Edc i.e. about 10% greater than in the case of a sine wave.
Consequently, if the problem of loss of controllability that was experienced with the conventional system of FIG. 1 can be overcome and a fundamental wave voltage can be output corresponding to single pulse mode, taking into account the fact that conventionally a margin of about 5% was applied, the voltage can be raised by 15%. The curves of voltage and flux in this case are indicated by the curves S2 and S2′ (in the case of single pulse operation) in FIG. 4.
Since a higher voltage can be achieved, the region of constant flux can be expanded up to a rotational speed that is 15% higher than conventionally, so the motor output capacity can be raised by 15% with exactly the same motor/inverter. Also, regarding the manner in which flux is weakened in the flux weakening region, weakening may be applied to a lesser degree. Since the torque generated by the motor is proportional to the product of the torque current and the flux-based current, if the flux is weakened the ratio of the current flowing in the motor to the torque is lowered. The fact that weakening may be applied to a lesser degree means that less current is required to generate the same torque i.e. efficiency can be improved.
With the system of FIG. 1, only sine wave PWM can be performed, so, rather than single-pulse current control, the current is indirectly controlled by changing over to another control system, not shown, such as phase control. If this is done, there is a large leap in the fundamental/lower-order harmonics of the voltage, so it is not possible to simply perform changeover but, instead, a complex changeover control process must be performed. Also, in the case of phase control, the current cannot be controlled at high speed, as in the case of current control.
With the system of FIG. 2, it was stated that a shift could be effected to single pulse operation without falling into a condition in which control could not be achieved in the high-speed region, but this is a manner of speaking focusing solely on the aspect of PWM control of current tracking PWM. Within the region in which sine wave PWM is possible, the current deviation is within the allowable error, but, outside the region in which sine wave PWM is possible, deviation increases. When the deviation increases, the voltage waveform approaches a single pulse. In the high-speed region in which sine wave PWM is impossible, the current deviation becomes so large that it is difficult to say that current control is being performed: however, rather than control becoming impossible, the current changes in response to changes in the current reference.